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19-4321; Rev 1; 12/09
KIT ATION EVALU BLE AVAILA
6A, 2MHz Step-Down Regulator with Integrated Switches
General Description Features
o Internal 26m RDS(ON) High-Side and 20m RDS(ON) Low-Side MOSFETs o Continuous 6A Output Current Over Temperature o 1% Output Accuracy Over Load, Line, and Temperature o Operates from 2.9V to 5.5V VIN Supply o Adjustable Output from 0.6V to (0.9 x VIN) o Soft-Start Reduces Inrush Supply Current o 500kHz to 2MHz Adjustable Switching Frequency o Compatible with Ceramic, Polymer, and Electrolytic Output Capacitors o Nine Preset and Adjustable Output Voltages 0.6V, 0.7V, 0.8V, 1.0V, 1.2V, 1.5V, 1.8V, 2.0V, 2.5V, and Adjustable o Monotonic Startup for Safe-Start Into Prebiased Outputs o Selectable Forced PWM or Skip Mode for Light Load Efficiency o Overcurrent and Overtemperature Protection o Output Current Sink/Source Capable with Cycleby-Cycle Protection o Open-Drain, Power-Good Output o Lead-Free, 4mm x 4mm, 24-Pin Thin QFN Package
MAX15039
The MAX15039 high-efficiency switching regulator delivers up to 6A load current at output voltages from 0.6V to 90% of VIN. The IC operates from 2.9V to 5.5V, making it ideal for on-board point-of-load and postregulation applications. Total output error is less than 1% over load, line, and temperature ranges. The MAX15039 features fixed-frequency PWM mode operation with a switching frequency range of 500kHz to 2MHz set by an external resistor. The MAX15039 provides the option of operating in a skip mode to improve light-load efficiency. High-frequency operation allows for an all-ceramic capacitor design. The high operating frequency also allows for small-size external components. The low-resistance on-chip nMOS switches ensure high efficiency at heavy loads while minimizing critical inductances, making the layout a much simpler task with respect to discrete solutions. Following a simple layout and footprint ensures first-pass success in new designs. The MAX15039 comes with a high bandwidth (28MHz) voltage-error amplifier. The voltage-mode control architecture and the voltage-error amplifier permit a type III compensation scheme to be utilized to achieve maximum loop bandwidth, up to 20% of the switching frequency. High loop bandwidth provides fast transient response, resulting in less required output capacitance and allowing for all-ceramic-capacitor designs. The MAX15039 provides two three-state logic inputs to select one of nine preset output voltages. The preset output voltages allow customers to achieve 1% output-voltage accuracy without using expensive 0.1% resistors. In addition, the output voltage can be set to any customer value by either using two external resistors at the feedback with a 0.6V internal reference or applying an external reference voltage to the REFIN input. The MAX15039 offers programmable soft-start time using one capacitor to reduce input inrush current.
Ordering Information
PART MAX15039ETG+ TEMP RANGE -40C to +85C PIN-PACKAGE 24 Thin QFN-EP*
+Denotes a lead(Pb)-free/RoHS-compliant package. *EP = Exposed pad.
Typical Operating Circuit
INPUT 2.9V TO 5.5V IN BST
EN VDD
MAX15039
LX OUT
OUTPUT 1.8V, 6A
Applications
Server Power Supplies POLs ASIC/CPU/DSP Core and I/O Voltages DDR Power Supplies Base-Station Power Supplies Telecom and Networking Power Supplies RAID Control Power Supplies
CTL2 CTL1 FREQ REFIN SS
PGND
FB
COMP
VDD
MODE GND PWRGD
Pin Configuration appears at end of data sheet.
________________________________________________________________ Maxim Integrated Products 1
For pricing, delivery, and ordering information, please contact Maxim Direct at 1-888-629-4642, or visit Maxim's website at www.maxim-ic.com.
6A, 2MHz Step-Down Regulator with Integrated Switches MAX15039
ABSOLUTE MAXIMUM RATINGS
IN, PWRGD to GND..................................................-0.3V to +6V VDD to GND ..................-0.3V to the lower of +4V or (VIN + 0.3V) COMP, FB, MODE, REFIN, CTL1, CTL2, SS, FREQ to GND ..........................................-0.3V to (VDD + 0.3V) OUT, EN to GND ......................................................-0.3V to +6V BST to LX..................................................................-0.3V to +6V BST to GND ............................................................-0.3V to +12V PGND to GND .......................................................-0.3V to +0.3V LX to PGND ..................-0.3V to the lower of +6V or (VIN + 0.3V) LX to PGND ..........-1V to the lower of +6V or (VIN + 1V) for 50ns ILX(RMS) (Note 1) ......................................................................6A VDD Output Short-Circuit Duration .............................Continuous Converter Output Short-Circuit Duration ....................Continuous Continuous Power Dissipation (TA = +70C) 24-Pin TQFN (derate 27.8mW/C above +70C) ........2222mW Thermal Resistance (Note 2) JA.................................................................................36C/W JC ..................................................................................6C/W Operating Temperature Range ...........................-40C to +85C Junction Temperature ......................................................+150C Storage Temperature Range .............................-65C to +150C Lead Temperature (soldering, 10s) .................................+300C
Note 1: LX has internal clamp diodes to PGND and IN. Applications that forward bias these diodes should take care not to exceed the IC's package power dissipation limits. Note 2: Package thermal resistances were obtained using the method described in JEDEC specification JESD51-7, using a fourlayer board. For detailed information on package thermal considerations, refer to www.maxim-ic.com/thermal-tutorial.
Stresses beyond those listed under "Absolute Maximum Ratings" may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability.
ELECTRICAL CHARACTERISTICS
(VIN = VEN = 5V, CVDD = 2.2F, TA = TJ = -40C to +85C, typical values are at TA = +25C, circuit of Figure 1, unless otherwise noted.) (Note 3)
PARAMETER IN IN Voltage Range IN Supply Current Total Shutdown Current from IN 3.3V LDO (VDD) VDD rising VDD Undervoltage Lockout Threshold VDD Output Voltage VDD Dropout VDD Current Limit BST BST Supply Current PWM COMPARATOR PWM Comparator Propagation Delay PWM Peak-to-Peak Ramp Amplitude PWM Valley Amplitude 10mV overdrive 20 1 0.8 ns V V VBST = VIN = 5V, VLX = 0 or 5V, VEN = 0V 0.025 A LX starts/stops switching VDD falling Minimum glitch-width rejection VIN = 5V, IVDD = 0 to 10mA VIN = 2.9V, IVDD = 10mA VIN = 5V, VDD = 0V 25 40 3.1 2.35 2.6 2.55 10 3.3 3.5 0.08 2.8 V s V V mA fS = 1MHz, no load VIN = 5V, VEN = 0V VIN = VDD = 3.3V, VEN = 0V VIN = 3.3V VIN = 5V 2.9 4.9 5.2 10 45 5.5 8 8.5 20 V mA A CONDITIONS MIN TYP MAX UNITS
2
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6A, 2MHz Step-Down Regulator with Integrated Switches
ELECTRICAL CHARACTERISTICS (continued)
(VIN = VEN = 5V, CVDD = 2.2F, TA = TJ = -40C to +85C, typical values are at TA = +25C, circuit of Figure 1, unless otherwise noted.) (Note 3)
PARAMETER ERROR AMPLIFIER COMP Clamp Voltage, High COMP Clamp Voltage, Low COMP Slew Rate COMP Shutdown Resistance Internally Preset Output Voltage Accuracy FB Set-Point Value FB to OUT Resistor Open-Loop Voltage Gain Error-Amplifier Unity-Gain Bandwidth Error-Amplifier Common-Mode Input Range Error-Amplifier Maximum Output Current FB Input Bias Current CTL_ CTL_ Input Bias Current VCTL_ = 0V VCTL_ = VDD Low, falling CTL_ Input Threshold Open High, rising Hysteresis REFIN REFIN Input Bias Current REFIN Offset Voltage LX (All Pins Combined) LX On-Resistance, High Side LX On-Resistance, Low Side ILX = -2A ILX = 2A High-side sourcing LX Current-Limit Threshold Low-side sinking Zero-crossing current threshold, MODE = VDD LX Leakage Current VIN = 5V, VEN = 0V VLX = 0V VLX = 5V VIN = VBST - VLX = 3.3V VIN = VBST - VLX = 5V VIN = 3.3V VIN = 5V 9 35 26 25 20 11 11 0.2 -0.01 -0.01 A A 35 45 m m VREFIN = 0.6V VREFIN = 0.9V, FB shorted to COMP -4.5 -185 +4.5 nA mV All VID transitions -7.2 7.2 0.8 VDD/2 VDD 0.8 50 mV V A VDD = 2.9V to 3.5V VCOMP = 1V, VREFIN = 0.6V VFB = 0.7V, sinking VFB = 0.5V, sourcing 0 1 -1 -125 VIN = 2.9V to 5V, VFB = 0.5V, VREFIN = 0.6V VIN = 2.9V to 5V, VFB = 0.7V, VREFIN = 0.6V VFB step from 0.5V to 0.7V in 10ns From COMP to GND, VIN = 3.3V, VCOMP = 100mV, VEN = VSS = 0V VREFIN = VSS, MODE = GND CTL1 = CTL2 = GND, MODE = GND All VID settings except CTL1 = CTL2 = GND -1 0.594 5.5 0.6 8 115 28 VDD - 2 2 0.7 1.6 6 +1 0.606 10.5 V V V/s % V k dB MHz V mA nA CONDITIONS MIN TYP MAX UNITS
MAX15039
CTL1 = CTL2 = GND
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3
6A, 2MHz Step-Down Regulator with Integrated Switches MAX15039
ELECTRICAL CHARACTERISTICS (continued)
(VIN = VEN = 5V, CVDD = 2.2F, TA = TJ = -40C to +85C, typical values are at TA = +25C, circuit of Figure 1, unless otherwise noted.) (Note 3)
PARAMETER LX Switching Frequency Switching Frequency Range LX Minimum Off-Time LX Maximum Duty Cycle LX Minimum Duty Cycle Average Short-Circuit IN Supply Current RMS LX Output Current ENABLE EN Input Logic-Low Threshold EN Input Logic-High Threshold EN Input Current MODE Logic-low, falling MODE Input-Logic Threshold MODE Input-Logic Hysteresis MODE Input Bias Current SS SS Current THERMAL SHUTDOWN Thermal-Shutdown Threshold Thermal-Shutdown Hysteresis POWER GOOD (PWRGD) Power-Good Threshold Voltage Power-Good Edge Deglitch PWRGD Output-Voltage Low PWRGD Leakage Current HICCUP OVERCURRENT LIMIT Current-Limit Startup Blanking Autoretry Restart Time 112 896 Clock Cycles Clock Cycles VFB falling, VREFIN = 0.6V VFB rising, VREFIN = 0.6V VFB rising or falling IPWRGD = 4mA VIN = VPWRGD = 5V, VFB = 0.7V, VREFIN = 0.6V 88 90 92.5 48 0.03 0.01 0.1 92 % VREFIN Clock Cycles V A Rising 165 25 C C VSS = 0.45V, VREFIN = 0.6V, sourcing 6.7 8 9.3 A Logic VDD/2 or open, rising Logic-high, rising MODE falling MODE = GND MODE = VDD 26 50 74 5 -5 5 %VDD A %VDD EN falling EN rising VEN = 0 or 5V, VIN = 5V 1.5 0.01 0.9 V V A RFREQ = 49.9k RFREQ = 49.9k OUT connected to GND, VIN = 5V 6 92 95 5 0.35 15 VIN = 2.9V to 5.5V CONDITIONS RFREQ = 49.9k RFREQ = 23.6k MIN 0.9 1.8 500 TYP 1 2 MAX 1.1 2.2 2000 78 UNITS MHz kHz ns % % A A
4
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6A, 2MHz Step-Down Regulator with Integrated Switches
ELECTRICAL CHARACTERISTICS (continued)
(VIN = VEN = 5V, CVDD = 2.2F, TA = TJ = -40C to +85C, typical values are at TA = +25C, circuit of Figure 1, unless otherwise noted.) (Note 3)
PARAMETER FB Hiccup Threshold Hiccup Threshold Blanking Time VFB falling VFB falling CONDITIONS MIN TYP 70 28 MAX UNITS % VREFIN s
MAX15039
Note 3: Specifications are 100% production tested at TA = +25C. Limits over the operating temperature range are guaranteed by design.
Typical Operating Characteristics
(Typical values are VIN = VEN = 5V, VOUT = 1.8V, RFREQ = 49.9k, IOUT = 6A, TA = +25C, circuit of Figure 1, unless otherwise noted.)
EFFICIENCY vs. OUTPUT CURRENT
MAX15039 toc01
EFFICIENCY vs. OUTPUT CURRENT
MAX15039 toc02
FREQUENCY vs. INPUT VOLTAGE
2.15 2.10 FREQUENCY (MHz)
MAX15039 toc03
100 90 EFFICIENCY (%) 80 70 60 VOUT = 1.2V 50 40 0.1 1.0 OUTPUT CURRENT (A) PWM SKIP
100 90 EFFICIENCY (%) 80 70 60 50 VIN = 3.3V 40 VOUT = 2.5V VOUT = 1.8V VOUT = 1.2V
2.20
VOUT = 2.5V VOUT = 1.8V
2.05 2.00 1.95 1.90 TA = +85C TA = +25C TA = -40C RFREQ = 23.2k 1.80 2.5 3.0 3.5 4.0 4.5 5.0 5.5 INPUT VOLTAGE (V)
PWM SKIP 1.0 OUTPUT CURRENT (A) 10.0
1.85
10.0
0.1
FREQUENCY vs. INPUT VOLTAGE
MAX15039 toc04
LOAD REGULATION
-0.05 OUTPUT-VOLTAGE CHANGE (%) -0.10 -0.15 -0.20 -0.25 -0.30 -0.35 -0.40 -0.45 -0.50 -0.12 0 1 2 3 4 5 6 7 2.5 5.5 VOUT = 1.8V VOUT = 2.5V VOUT = 1.2V
MAX15039 toc05a
LINE REGULATION (LOAD = 6A)
MAX15039 toc05b
1.20 1.15 1.10 FREQUENCY (MHz) 1.05 1.00 0.95 0.90 0.85 0.80 2.5 3.0 3.5 4.0 4.5 5.0 INPUT VOLTAGE (V) TA = -40C RFREQ = 49.9k TA = +25C TA = +85C
0
0 OUTPUT-VOLTAGE CHANGE (%) -0.02 VOUT = 1.8V -0.04 -0.06 -0.08 VOUT = 1.2V -0.10
3.0
3.5
4.0
4.5
5.0
5.5
LOAD CURRENT (A)
INPUT VOLTAGE (V)
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5
6A, 2MHz Step-Down Regulator with Integrated Switches MAX15039
Typical Operating Characteristics (continued)
(Typical values are VIN = VEN = 5V, VOUT = 1.8V, RFREQ = 49.9k, IOUT = 6A, TA = +25C, circuit of Figure 1, unless otherwise noted.)
LOAD TRANSIENT
MAX15039 toc06
SWITCHING WAVEFORMS (FORCED PWM, 2A LOAD)
MAX15039 toc07
SWITCHING WAVEFORMS (SKIP MODE, NO LOAD)
MAX15039 toc08
VOUT AC-COUPLED VOUT 100mV/div
VOUT AC-COUPLED 50mV/div
AC-COUPLED 100mV/div
1A/div 2A/div 2A IOUT 0A ILX 0A 5V/div VLX 5V/div VLX 0V ILX 0A
40s/div
400ns/div
2s/div
SOFT-START WAVEFORM (RLOAD = 0.5)
MAX15039 toc09
SHUTDOWN WAVEFORM (RLOAD = 0.5)
MAX15039 toc10
VEN 5V/div
VEN 5V/div
VOUT 1V/div 0V
VOUT 1V/div 0V
400s/div
10s/div
INPUT SHUTDOWN CURRENT vs. INPUT VOLTAGE
MAX15039 toc11
MAXIMUM OUTPUT CURRENT vs. OUTPUT VOLTAGE
9 8 7 6 5 4 3
MAX15039 toc12
12 INPUT SHUTDOWN CURRENT (A) 11 10 9 8 7 6 VEN = 0V 5 2.5 3.0 3.5 4.0 4.5 5.0
10 MAXIMUM OUTPUT CURRENT (A)
2 5.5 0.5 1.0 1.5 2.0 2.5 INPUT VOLTAGE (V) OUTPUT VOLTAGE (V)
6
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6A, 2MHz Step-Down Regulator with Integrated Switches
Typical Operating Characteristics (continued)
(Typical values are VIN = VEN = 5V, VOUT = 1.8V, RFREQ = 49.9k, IOUT = 6A, TA = +25C, circuit of Figure 1, unless otherwise noted.)
MAX15039
HICCUP CURRENT LIMIT
MAX15039 toc13
RMS INPUT CURRENT DURING SHORT CIRCUIT vs. INPUT VOLTAGE
MAX15039 toc14
EXPOSED PAD TEMPERATURE vs. AMBIENT TEMPERATURE
90 EXPOSED PAD TEMPERATURE (C) 80 70 60 50 40 30 20 10 0 MEASURED ON A MAX15039EVKIT 0 20 40 60 80 100 6A LOAD
MAX15039 toc15 MAX15039 toc17
0.8 1V/div RMS INPUT CURRENT (A) 0V 0.7 0.6 0.5 0.4 0.3 0.2 0.1 VOUT = 0V 0 2.5 3.0 3.5 4.0 4.5 5.0
100
VOUT
IOUT
5A/div 0A
IIN
1A/div 0A
400s/div
5.5
INPUT VOLTAGE (V)
AMBIENT TEMPERATURE (C)
FEEDBACK VOLTAGE vs. TEMPERATURE
0.63 FEEDBACK VOLTAGE (V) 0.62 0.61 0.60 0.59 0.58 0.57 0.56 -40 -15 10 35 60 85 VPWRGD VOUT VREFIN
MAX15039 toc16
SOFT-START WITH REFIN
1A/div 0A 0.5V/div 0V 1V/div 0V 2V/div 0V 200s/div
0.64
IIN
TEMPERATURE (C)
STARTING INTO PREBIASED OUTPUT (MODE = VDD, VOUT = 2.5V, 2A LOAD)
MAX15039 toc18
STARTING INTO PREBIASED OUTPUT (MODE = VDD/2, VOUT = 2.5V, 2A LOAD)
MAX15039 toc19
VEN
5V/div 0V 1V/div
VEN
5V/div 0V 1V/div
VOUT 0V 2A IOUT 0A VPWRGD 200s/div 5V/div 0V
VOUT 0V 2A IOUT 0A VPWRGD 200s/div 5V/div 0V
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7
6A, 2MHz Step-Down Regulator with Integrated Switches MAX15039
Typical Operating Characteristics (continued)
(Typical values are VIN = VEN = 5V, VOUT = 1.8V, RFREQ = 49.9k, IOUT = 6A, TA = +25C, circuit of Figure 1, unless otherwise noted.)
STARTING INTO PREBIASED OUTPUT (MODE = VDD, VOUT = 2.5V, NO LOAD)
MAX15039 toc20
STARTING INTO PREBIASED OUTPUT (MODE = VDD/2, VOUT = 2.5V, NO LOAD)
MAX15039 toc21
VEN 2V/div 0V VOUT 1V/div 0V VPWRGD 2V/div 0V
VEN 2V/div 0V VOUT 1V/div 0V VPWRGD 2V/div 0V
200s/div
200s/div
STARTING INTO PREBIASED OUTPUT ABOVE NOMINAL SET POINT (VOUT = 1.5V)
MAX15039 toc22
STARTING INTO PREBIASED ABOVE NOMINAL SET POINT (VOUT = 1.5V)
MAX15039 toc23
VEN 2V/div 0V VOUT 1V/div 0V VPWRGD 2V/div VMODE = VDD, NO LOAD 1ms/div 0V
VEN 2V/div 0V VOUT 1V/div 0V VPWRGD 2V/div 0V
VOUT = 1.5V, VMODE = VDD/2, NO LOAD 1ms/div
TRANSITION FROM SKIP MODE TO FORCED PWM MODE
MAX15039 toc24
TRANSITION FROM FORCED PWM TO SKIP MODE
MAX15039 toc25
VMODE 5V/div VLX 5V/div VOUT 0.5V/div
VMODE 5V/div VLX 5V/div VOUT 0.5V/div
NO LOAD 0V 2ms/div 4ms/div
NO LOAD
0V
8
_______________________________________________________________________________________
6A, 2MHz Step-Down Regulator with Integrated Switches
Pin Description
PIN 1 2 3 4 5 NAME MODE VDD CTL1 CTL2 REFIN FUNCTION Functional Mode Selection Input. See the MODE Selection section for more information. 3.3V LDO Output. Supply input for the internal analog core. Connect a low-ESR, ceramic capacitor with a minimum value of 2.2F from VDD to GND. Preset Output-Voltage Selection Inputs. CTL1 and CTL2 set the output voltage to one of nine preset voltages. See Table 1 and the Programming the Output Voltage (CTL1, CTL2) section for preset voltages. External Reference Input. Connect REFIN to SS to use the internal 0.6V reference. Connecting REFIN to an external voltage forces FB to regulate to the voltage applied to REFIN. REFIN is internally pulled to GND when the IC is in shutdown/hiccup mode. Soft-Start Input. Connect a capacitor from SS to GND to set the startup time. Use a capacitor with a 1nF minimum value. See the Soft-Start and REFIN section for details on setting the soft-start time. Analog Ground Connection. Connect GND and PGND together at one point near the input bypass capacitor return terminal. Voltage Error-Amplifier Output. Connect the necessary compensation network from COMP to FB and OUT. COMP is internally pulled to GND when the IC is in shutdown/hiccup mode. Feedback Input. Connect FB to the center tap of an external resistive divider from the output to GND to set the output voltage from 0.6V to 90% of VIN. Connect FB through an RC network to the output when using CTL1 and CTL2 to select any of nine preset voltages. Output-Voltage Sense. Connect to the converter output. Leave OUT unconnected when an external resistive divider is used. Oscillator Frequency Select. Connect a precision resistor from FREQ to GND to select the switching frequency. See the Frequency Select (FREQ) section. Open-Drain, Power-Good Output. PWRGD is high impedance when VFB rises above 92.5% (typ) of VREFIN and VREFIN is above 0.54V. PWRGD is internally pulled low when VFB falls below 90% (typ) of VREFIN or VREFIN is below 0.54V. PWRGD is internally pulled low when the IC is in shutdown mode, VDD is below the internal UVLO threshold, or the IC is in thermal shutdown. High-Side MOSFET Driver Supply. Internally connected to IN through a pMOS switch. Bypass BST to LX with a 0.1F capacitor. Inductor Connection. All LX pins are internally shorted together. Connect all LX pins to the switched side of the inductor. LX is high impedance when the IC is in shutdown mode. Power Ground. Connect all PGND pins externally to the power ground plane. Connect all PGND pins together near the IC. Input Power Supply. Input supply range is from 2.9V to 5.5V. Bypass IN to PGND with a 22F ceramic capacitor. Enable Input. Logic input to enable/disable the MAX15039. Exposed Pad. Solder EP to a large contiguous copper plane connected to PGND to optimize thermal performance. Do not use EP as a ground connection for the device.
MAX15039
6 7 8
SS GND COMP
9
FB
10 11
OUT FREQ
12
PWRGD
13 14, 15, 16 17-20 21, 22, 23 24 --
BST LX PGND IN EN EP
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9
6A, 2MHz Step-Down Regulator with Integrated Switches MAX15039
Block Diagram
VDD
3.3V LDO EN SHUTDOWN CONTROL UVLO CIRCUITRY
MAX15039
BST CURRENT-LIMIT COMPARATOR BIAS GENERATOR BST SWITCH IN
VOLTAGE REFERENCE
THERMAL SHUTDOWN
CONTROL LOGIC IN
LX
SS
SOFT-START PGND CURRENT-LIMIT COMPARATOR
REFIN OUT 8k FB CTL1 CTL2 VID VOLTAGECONTROL CIRCUITRY ERROR AMPLIFIER PWM COMPARATOR
MODE
1VP-P OSCILLATOR
FREQ
COMP PWRGD SHDN FB COMP CLAMPS 0.9 x VREFIN GND
10
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6A, 2MHz Step-Down Regulator with Integrated Switches MAX15039
INPUT 2.9V TO 5.5V IN C6 22F C7 0.1F BST C10 0.1F L1 0.47H LX VDD C5 2.2F CTL2 CTL1 PGND EN FB FREQ REFIN R4 49.9k SS C4 0.022F COMP MODE GND PWRGD C1 33pF VDD R1 20k C2 1500pF R2 2.67k OUT C3 560pF R3 158 C8 22F C9 0.01F 2.2 OPTIONAL C15 1000pF OUTPUT 1.8V, 6A
MAX15039
Figure 1. Typical Application Circuit: 1MHz, All-Ceramic-Capacitor Design with VIN = 2.9V to 5.5V and VOUT = 1.8V
Detailed Description
The MAX15039 high-efficiency, voltage-mode switching regulator delivers up to 6A of output current. The MAX15039 provides output voltages from 0.6V to 0.9 x VIN from 2.9V to 5.5V input supplies, making it ideal for on-board point-of-load applications. The output-voltage accuracy is better than 1% over load, line, and temperature. The MAX15039 features a wide switching frequency range, allowing the user to achieve all-ceramic-capacitor designs and fast transient responses (see Figure 1). The high operating frequency minimizes the size of external components. The MAX15039 is available in a small (4mm x 4mm), lead-free, 24-pin thin QFN package. The REFIN function makes the MAX15039 an ideal candidate for DDR and tracking power supplies. Using internal low-RDS(ON) (20m for the low-side n-channel
MOSFET and 26m for the high-side n-channel MOSFET) maintains high efficiency at both heavy-load and high-switching frequencies. The MAX15039 employs voltage-mode control architecture with a high bandwidth (28MHz) error amplifier. The voltage-mode control architecture allows up to 2MHz switching frequency, reducing board area. The op amp voltage-error amplifier works with type III compensation to fully utilize the bandwidth of the high-frequency switching to obtain fast transient response. Adjustable soft-start time provides flexibilities to minimize input startup inrush current. An open-drain, power-good (PWRGD) output goes high when VFB reaches 92.5% of VREFIN and VREFIN is greater than 0.54V. The MAX15039 provides an option for three modes of operation: regular PWM, PWM mode with monotonic startup into prebiased output, or skip mode with monotonic startup into prebiased output.
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11
6A, 2MHz Step-Down Regulator with Integrated Switches MAX15039
Controller Function
The controller logic block is the central processor that determines the duty cycle of the high-side MOSFET under different line, load, and temperature conditions. Under normal operation, where the current-limit and temperature protection are not triggered, the controller logic block takes the output from the PWM comparator and generates the driver signals for both high-side and low-side MOSFETs. The break-before-make logic and the timing for charging the bootstrap capacitors are calculated by the controller logic block. The error signal from the voltage-error amplifier is compared with the ramp signal generated by the oscillator at the PWM comparator and, thus, the required PWM signal is produced. The high-side switch is turned on at the beginning of the oscillator cycle and turns off when the ramp voltage exceeds the VCOMP signal or the current-limit threshold is exceeded. The low-side switch is then turned on for the remainder of the oscillator cycle. C= 8 A x t SS 0 . 6V
where tSS is the required soft-start time in seconds. The MAX15039 also features an external reference input (REFIN). The IC regulates FB to the voltage applied to REFIN. The internal soft-start is not available when using an external reference. A method of soft-start when using an external reference is shown in Figure 2. Connect REFIN to SS to use the internal 0.6V reference. Use a capacitor of 1nF minimum value at SS.
Undervoltage Lockout (UVLO)
The UVLO circuitry inhibits switching when VDD is below 2.55V (typ). Once VDD rises above 2.6V (typ), UVLO clears and the soft-start function activates. A 50mV hysteresis is built in for glitch immunity.
BST
The gate-drive voltage for the high-side, n-channel switch is generated by a flying-capacitor boost circuit. The capacitor between BST and LX is charged from the VIN supply while the low-side MOSFET is on. When the low-side MOSFET is switched off, the voltage of the capacitor is stacked above LX to provide the necessary turn-on voltage for the high-side internal MOSFET.
Current Limit
The internal, high-side MOSFET has a typical 11A peak current-limit threshold. When current flowing out of LX exceeds this limit, the high-side MOSFET turns off and the synchronous rectifier turns on. The synchronous rectifier remains on until the inductor current falls below the low-side current limit. This lowers the duty cycle and causes the output voltage to droop until the current limit is no longer exceeded. The MAX15039 uses a hiccup mode to prevent overheating during short-circuit output conditions. During current limit, if VFB drops below 70% of VREFIN and stays below this level for 12s or more, the MAX15039 enters hiccup mode. The high-side MOSFET and the synchronous rectifier are turned off and both COMP and REFIN are internally pulled low. If REFIN and SS are connected together, both are pulled low. The part remains in this state for 896 clock cycles and then attempts to restart for 112 clock cycles. If the fault causing current limit has cleared, the part resumes normal operation. Otherwise, the part reenters hiccup mode again.
Frequency Select (FREQ)
The switching frequency is resistor programmable from 500kHz to 2MHz. Set the switching frequency of the IC with a resistor (RFREQ) connected from FREQ to GND. RFREQ is calculated as: RFREQ = 50k 1 x( 0.95s fS
- 0.05s)
where fS is the desired switching frequency in Hertz.
R1 REFIN
Soft-Start and REFIN
The MAX15039 utilizes an adjustable soft-start function to limit inrush current during startup. An 8A (typ) current source charges an external capacitor connected to SS. The soft-start time is adjusted by the value of the external capacitor from SS to GND. The required capacitance value is determined as:
R2 C
MAX15039
Figure 2. Typical Soft-Start Implementation with External Reference
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6A, 2MHz Step-Down Regulator with Integrated Switches
Power-Good Output (PWRGD)
PWRGD is an open-drain output that goes high impedance when VFB is above 0.925 x VREFIN and VREFIN is above 0.54V for at least 48 clock cycles. PWRGD pulls low when V FB is below 90% of V REFIN or V REFIN is below 0.54V for at least 48 clock cycles. PWRGD is low when the IC is in shutdown mode, VDD is below the internal UVLO threshold, or the IC is in thermal shutdown mode.
Table 1. CTL1 and CTL2 Output Voltage Selection
CTL1 GND VDD GND GND Unconnected Unconnected Unconnected VDD VDD CTL2 GND VDD Unconnected VDD GND Unconnected VDD GND Unconnected VOUT (V) 0.6 0.7 0.8 1.0 1.2 1.5 1.8 2.0 2.5
MAX15039
Programming the Output Voltage (CTL1, CTL2)
As shown in Table 1, the output voltage is pin programmable by the logic states of CTL1 and CTL2. CTL1 and CTL2 are trilevel inputs: VDD, unconnected, and GND. An 8.06k resistor must be connected between OUT and FB when CTL1 and CTL2 are connected to GND. The logic states of CTL1 and CTL2 should be programmed only before power-up. Once the part is enabled, CTL1 and CTL2 should not be changed. If the output voltage needs to be reprogrammed, cycle power or EN and reprogram before enabling. The output voltage can be programmed continuously from 0.6V to 90% of VIN by using a resistor-divider network from VOUT to FB to GND as shown in Figure 3a. CTL1 and CTL2 must be connected to GND.
L LX COUT
VOUT
MAX15039
OUT FB R1 CTL2 COMP C2 C1
R3
R2 C3
CTL1
Shutdown Mode
Drive EN to GND to shut down the IC and reduce quiescent current to 10A (typ). During shutdown, the LX is high impedance. Drive EN high to enable the MAX15039.
R4
a) EXTERNAL RESISTIVE DIVIDER
Thermal Protection
Thermal-overload protection limits total power dissipation in the device. When the junction temperature exceeds TJ = +165C, a thermal sensor forces the device into shutdown, allowing the die to cool. The thermal sensor turns the device on again after the junction temperature cools by 20C, causing a pulsed output during continuous overload conditions. The soft-start sequence begins after recovery from a thermal-shutdown condition.
LX
L COUT
VOUT
MAX15039
OUT
R2
R3 8k FB VOLTAGE SELECT CTL1 CTL2 COMP C2
C3
Applications Information
IN and VDD Decoupling
To decrease the noise effects due to the high switching frequency and maximize the output accuracy of the MAX15039, decouple IN with a 22F capacitor from IN to PGND. Also decouple VDD with a 2.2F low-ESR ceramic capacitor from V DD to GND. Place these capacitors as close as possible to the IC.
R1
C1
b) INTERNAL PRESET VOLTAGES
Figure 3. Type III Compensation Network
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13
6A, 2MHz Step-Down Regulator with Integrated Switches MAX15039
Inductor Selection
Choose an inductor with the following equation: L= VOUT x (VIN - VOUT ) fS x VIN x LIR x IOUT(MAX) evaluation circuit. A smaller ripple current results in less output-voltage ripple. Since the inductor ripple current is a factor of the inductor value, the output-voltage ripple decreases with larger inductance. Use ceramic capacitors for low ESR and low ESL at the switching frequency of the converter. The ripple voltage due to ESL is negligible when using ceramic capacitors. Load-transient response depends on the selected output capacitance. During a load transient, the output instantly changes by ESR x ILOAD. Before the controller can respond, the output deviates further, depending on the inductor and output capacitor values. After a short time, the controller responds by regulating the output voltage back to its predetermined value. The controller response time depends on the closed-loop bandwidth. A higher bandwidth yields a faster response time, preventing the output from deviating further from its regulating value. See the Compensation Design section for more details.
where LIR is the ratio of the inductor ripple current to full load current at the minimum duty cycle. Choose LIR between 20% to 40% for best performance and stability. Use an inductor with the lowest possible DC resistance that fits in the allotted dimensions. Powdered iron ferrite core types are often the best choice for performance. With any core material, the core must be large enough not to saturate at the current limit of the MAX15039.
Output-Capacitor Selection
The key selection parameters for the output capacitor are capacitance, ESR, ESL, and voltage-rating requirements. These affect the overall stability, output ripple voltage, and transient response of the DC-DC converter. The output ripple occurs due to variations in the charge stored in the output capacitor, the voltage drop due to the capacitor's ESR, and the voltage drop due to the capacitor's ESL. Estimate the output-voltage ripple due to the output capacitance, ESR, and ESL: VRIPPLE = VRIPPLE(C) + VRIPPLE(ESR) + VRIPPLE(ESL) where the output ripple due to output capacitance, ESR, and ESL is: IP -P VRIPPLE(C) = 8 x C OUT x fS VRIPPLE(ESR) = IP -P x ESR I VRIPPLE(ESL) = P -P x ESL t ON or: IP VRIPPLE(ESL) = P -P x ESL t OFF
Input-Capacitor Selection
The input capacitor reduces the current peaks drawn from the input power supply and reduces switching noise in the IC. The total input capacitance must be equal or greater than the value given by the following equation to keep the input-ripple voltage within specification and minimize the high-frequency ripple current being fed back to the input source: CIN _ MIN = D x TS x IOUT VIN - RIPPLE
or whichever is larger. The peak-to-peak inductor current (IP-P) is: V -V V IP - P = IN OUT x OUT fS x L VIN Use these equations for initial output-capacitor selection. Determine final values by testing a prototype or an
where VIN-RIPPLE is the maximum allowed input ripple voltage across the input capacitors and is recommended to be less than 2% of the minimum input voltage. D is the duty cycle (VOUT/VIN) and TS is the switching period (1/fS). The impedance of the input capacitor at the switching frequency should be less than that of the input source so high-frequency switching currents do not pass through the input source, but are instead shunted through the input capacitor. The input capacitor must meet the ripple current requirement imposed by the switching currents. The RMS input ripple current is given by: IRIPPLE = ILOAD x VOUT x (VIN - VOUT ) VIN
where IRIPPLE is the input RMS ripple current.
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6A, 2MHz Step-Down Regulator with Integrated Switches
Compensation Design
The power transfer function consists of one double pole and one zero. The double pole is introduced by the inductor L and the output capacitor CO. The ESR of the output capacitor determines the zero. The double pole and zero frequencies are given as follows: fP1_ LC = fP2 _ LC = 1 R + ESR 2 x L x C O x O R O + RL 1 2 x ESR x C O
MAX15039
fP3 _ EA = fP2 _ EA =
1 2 x R1 x C2 1 2 x R2 x C3
f Z _ ESR =
where RL is equal to the sum of the output inductor's DCR (DC resistance) and the internal switch resistance, RDS(ON). A typical value for RDS(ON) is 20m (low-side MOSFET) and 26m (high-side MOSFET). RO is the output load resistance, which is equal to the rated output voltage divided by the rated output current. ESR is the total equivalent series resistance of the output capacitor. If there is more than one output capacitor of the same type in parallel, the value of the ESR in the above equation is equal to that of the ESR of a single output capacitor divided by the total number of output capacitors. The high switching frequency range of the MAX15039 allows the use of ceramic output capacitors. Since the ESR of ceramic capacitors is typically very low, the frequency of the associated transfer function zero is higher than the unity-gain crossover frequency, fC, and the zero cannot be used to compensate for the double pole created by the output filtering inductor and capacitor. The double pole produces a gain drop of 40dB/decade and a phase shift of 180. The compensation network error amplifier must compensate for this gain drop and phase shift to achieve a stable high-bandwidth closed-loop system. Therefore, use type III compensation as shown in Figures 3 and 4. Type III compensation possesses three poles and two zeros with the first pole, fP1_EA, located at zero frequency (DC). Locations of other poles and zeros of the type III compensation are given by: fZ1_ EA = fZ2 _ EA = 1 2 x R1 x C1 1 2 x R3 x C3
The above equations are based on the assumptions that C1 >> C2 and R3 >> R2, which are true in most applications. Placements of these poles and zeros are determined by the frequencies of the double pole and ESR zero of the power transfer function. It is also a function of the desired close-loop bandwidth. The following section outlines the step-by-step design procedure to calculate the required compensation components for the MAX15039. When the output voltage of the MAX15039 is programmed to a preset voltage, R3 is internal to the IC and R4 does not exist (Figure 3b). When externally programming the MAX15039 (Figure 3a), the output voltage is determined by: R4 = 0.6 x R3 (for VOUT > 0.6V) (VOUT - 0.6)
For a 0.6V output, connect an 8.06k resistor from FB to OUT. The zero-cross frequency of the close-loop, fC, should be between 10% and 20% of the switching frequency, fS. A higher zero-cross frequency results in faster transient response. Once fC is chosen, C1 is calculated from the following equation: VIN VP - P C1 = R 2 x x R3 x (1 + L ) x fC RO 2.5 x where VP-P is the ramp peak-to-peak voltage (1V typ). Due to the underdamped nature of the output LC double pole, set the two zero frequencies of the type III compensation less than the LC double-pole frequency to provide adequate phase boost. Set the two zero frequencies to 80% of the LC double-pole frequency. Hence: R1 = 1 x 0. 8 x C1 L x C O x (R O + ESR) RL + R O
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15
6A, 2MHz Step-Down Regulator with Integrated Switches MAX15039
C3 = 1 x 0. 8 x R3 L x C O x (R O + ESR) RL + R O
Table 2. Mode Selection
MODE CONNECTION GND Unconnected or VDD/2 VDD OPERATION MODE Forced PWM Forced PWM. Soft-start up into a prebiased output (monotonic startup). Skip Mode. Soft-start into a prebiased output (monotonic startup).
Setting the second compensation pole, f P2_EA , at fZ_ESR yields: C x ESR R2 = O C3 Set the third compensation pole at 1/2 of the switching frequency. Calculate C2 as follows: 1 C2 = x R1 x fS The above equations provide application compensation when the zero-cross frequency is significantly higher than the double-pole frequency. When the zero-cross frequency is near the double-pole frequency, the actual zerocross frequency is higher than the calculated frequency. In this case, lowering the value of R1 reduces the zerocross frequency. Also, set the third pole of the type III compensation close to the switching frequency if the zero-cross frequency is above 200kHz to boost the phase margin. The recommended range for R3 is 2k to 10k. Note that the loop compensation remains unchanged if only R4's resistance is altered to set different outputs.
COMPENSATION TRANSFER FUNCTION DOUBLE POLE GAIN (dB) POWER-STAGE TRANSFER FUNCTION
OPEN-LOOP GAIN THIRD POLE
SECOND POLE
FIRST AND SECOND ZEROS
MODE Selection
The MAX15039 features a mode selection input (MODE) that users can select a functional mode for the device (see Table 2).
Figure 4. Type III Compensation Illustration
Soft-Starting Into a Prebiased Output Mode (Monotonic Startup)
When MODE is left unconnected or biased to VDD/2, the MAX15039 soft-starts into a prebiased output without discharging the output capacitor. This type of operation is also termed monotonic startup. See the Starting Into Prebiased Output waveforms in the Typical Operating Characteristics section for an example. In monotonic startup mode, both low-side and highside switches remain off to avoid discharging the prebiased output. PWM operation starts when the FB voltage crosses the SS voltage. As in forced-PWM mode, the PWM activity starts with the low-side switch turning on first to build the bootstrap capacitor charge. The MAX15039 is also able to start into prebiased with the output above the nominal set point without abruptly discharging the output, thanks to the sink current control of the low-side switch through a 4-step DAC in 128 clock cycles. Monotonic startup mode automatically switches to forced-PWM mode 4096 clock cycles delay after the voltage at FB increases above 92.5% of VREFIN. The additional delay prevents an early transi-
Forced-PWM Mode
Connect MODE to GND to select forced-PWM mode. In forced-PWM mode, the MAX15039 operates at a constant switching frequency (set by the resistor at FREQ terminal) with no pulse skipping. PWM operation starts after a brief settling time when EN goes high. The lowside switch turns on first, charging the bootstrap capacitor to provide the gate-drive voltage for the highside switch. The low-side switch turns off either at the end of the clock period or once the low-side switch sinks 1.35A current (typ), whichever occurs first. If the low-side switch is turned off before the end of the clock period, the high-side switch is turned on for the remaining part of the time interval until the inductor current reaches 0.9A, or the end of clock cycle is encountered. Starting from the first PWM activity, the sink current threshold is increased through an internal 4-step DAC to reach the current limit of 11A after 128 clock periods. This is done to help a smooth recovery of the regulated voltage even in case of accidental prebiased output in spite of the initial forced-PWM mode selection.
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6A, 2MHz Step-Down Regulator with Integrated Switches
tion from monotonic startup to forced-PWM mode during soft-start when a prolonged time constant external REFIN voltage is applied. The maximum allowed soft-start time is 2ms when an external reference is applied at REFIN in the case of starting up into prebiased output.
PCB Layout Considerations and Thermal Performance
Careful PCB layout is critical to achieve clean and stable operation. It is highly recommended to duplicate the MAX15039 EV kit layout for optimum performance. If deviation is necessary, follow these guidelines for good PCB layout: 1) Connect input and output capacitors to the power ground plane; connect all other capacitors to the signal ground plane. 2) Place capacitors on VDD, IN, and SS as close as possible to the IC and its corresponding pin using direct traces. Keep power ground plane (connected to PGND) and signal ground plane (connected to GND) separate. 3) Keep the high-current paths as short and wide as possible. Keep the path of switching current short and minimize the loop area formed by LX, the output capacitors, and the input capacitors. 4) Connect IN, LX, and PGND separately to a large copper area to help cool the IC to further improve efficiency and long-term reliability. 5) Ensure all feedback connections are short and direct. Place the feedback resistors and compensation components as close as possible to the IC. 6) Route high-speed switching nodes, such as LX, away from sensitive analog areas (FB, COMP).
MAX15039
Skip Mode
Connect MODE to VDD to select skip mode. In skip mode, the MAX15039 switches only as necessary to maintain the output at light loads (not capable of sinking current from the output), but still operates with fixed-frequency (set by the resistor at FREQ terminal) PWM at medium and heavy loads. This maximizes light-load efficiency and reduces the input quiescent current. In case of prolonged high-side idle activity (beyond eight clock cycles), the low-side switch is turned on briefly to rebuild the charge lost in the bootstrap capacitor before the next on-cycle of the high-side switch. In skip mode, the low-side switch is turned off when the inductor current decreases to 0.2A (typ) to ensure no reverse current flowing from the output capacitor and the best conversion efficiency/minimum supply current. The high-side switch minimum on-time is controlled to guarantee that 0.9A current is reached to avoid high frequency bursts at no load conditions and that might cause a rapid increase of the supply current caused by additional switching losses. Even if skip mode is selected at the device turn-on, the monotonic startup mode is internally selected during soft-start. The transition to skip mode is automatically achieved 4096 clock cycles after the voltage at FB increases above 92.5% of VREFIN. Changing from skip mode to forced-PWM mode and vice-versa can be done at any time. The output capacitor should be large enough to limit the output-voltage overshoot/undershoot due to the settling times to reach different duty-cycle set points corresponding to forcedPWM mode and skip mode at light loads.
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17
6A, 2MHz Step-Down Regulator with Integrated Switches MAX15039
Pin Configuration
PROCESS: BiCMOS
PGND PGND
Chip Information
TOP VIEW
18 PGND 19 PGND 20 IN 21 IN 22 IN 23 EN 24
17
16
15
14
13 12 PWRGD 11 FREQ 10 OUT
BST
LX
LX
LX
Package Information
For the latest package outline information and land patterns, go to www.maxim-ic.com/packages. PACKAGE TYPE 24 TQFN-EP PACKAGE CODE T2444-4 DOCUMENT NO. 21-0139
MAX15039
*EP
9 8
FB COMP GND
+
1 MODE 2 VDD 3 CTL1 4 CTL2 5 REFIN 6 SS
7
THIN QFN
*EXPOSED PAD
18
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6A, 2MHz Step-Down Regulator with Integrated Switches
Revision History
REVISION NUMBER 0 1 REVISION DATE 10/08 12/09 Initial release Updated the Typical Operating Characteristics. DESCRIPTION PAGES CHANGED -- 5
MAX15039
Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are implied. Maxim reserves the right to change the circuitry and specifications without notice at any time.
Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600 ____________________ 19
(c) 2009 Maxim Integrated Products Maxim is a registered trademark of Maxim Integrated Products, Inc.

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